1. Field of the Invention
The present invention relates to electronic ballasts and, more particularly, to electronic dimming ballasts for gas discharge lamps, such as fluorescent lamps.
2. Description of the Related Art
Electronic ballasts for fluorescent lamps typically include a “front end” and a “back end”. The front end typically includes a rectifier for changing alternating-current (AC) mains line voltage to a direct-current (DC) bus voltage and a filter circuit for filtering the DC bus voltage. The ballast back end typically includes a switching inverter for converting the DC bus voltage to a high-frequency AC voltage, and a resonant tank circuit having a relatively high output impedance for coupling the high-frequency AC voltage to the lamp electrodes.
The front end of electronic ballasts also often include a boost converter, which is an active circuit for boosting the magnitude of the DC bus voltage above peak of line voltage, and for improving the total harmonic distortion (THD) and power factor of the input current to the ballast. However, boost converters typically include integrated circuits (IC) and semiconductor switches, such as field effect transistors (FETs). In order to handle the amount of current required to drive the lamp at high end (i.e. at or near 100% light intensity), the components of such a boost converter are typically large and costly.
A prior art ballast 100 will be described with reference to the block diagram shown in FIG. 1 and the voltage and current waveforms shown in FIGS. 2a-2d and is explained in greater detail in U.S. Pat. No. 6,674,248, issued on Jan. 6, 2004, entitled “Electronic Ballast”, which is herein incorporated by reference in its entirety.
The ballast 100 includes an electromagnetic interference (EMI) filter 115 and a rectifier 120 both capable of being connected to an AC power supply such as a typical 120V, 60 Hz AC main. The EMI filter 115 isolates high-frequency noise generated by the ballast circuitry from the AC power supply. The rectifier 120 converts the AC input voltage to a rectified pulsating DC voltage 210, which has a maximum value of VPEAK (shown as 230 in FIG. 2a). For example, if the AC input voltage has an RMS (root mean square) value of 277V, the value of VPEAK will be approximately 392V. The rectifier 120 is connected to a valley-fill circuit 130 through a diode 140. A high-frequency filter capacitor 150 is connected across the inputs to the valley-fill circuit 130. The valley-fill circuit 130 selectively charges and discharges an energy-storage device, such as one or more capacitors, so as to fill the “valleys” between successive rectified voltage peaks to produce a substantially DC bus voltage 220. The DC bus voltage is the greater of either the rectified voltage, or the voltage across the energy storage device in the valley-fill circuit 130.
The outputs of the valley-fill circuit 130 are in turn connected to the inputs to an inverter 160. The inverter 160 converts the rectified DC voltage to a high-frequency AC voltage. The outputs of the inverter 160 are connected to an output circuit 170, which typically includes a resonant tank, and may also include a coupling transformer. The output circuit filters the inverter 160 output to supply essentially sinusoidal voltage, as well as provide voltage gain and increased output impedance. The output circuit 170 is capable of being connected to drive a load 180 such as a gas discharge lamp; for example, a fluorescent lamp.
An output current sense circuit 185 coupled to the load 180 provides load current feedback to a control circuit 190. The control circuit 190 generates control signals to control the operation of the valley-fill circuit 130 and the inverter 160 so as to provide a desired load current to the load 180. A power supply 110 is connected across the outputs of the rectifier 120 to provide the necessary power for proper operation of the control circuit 190.
A schematic representation of a prior art valley-fill circuit 330 that may be used with ballast 100 is shown in FIG. 3a. The rectified pulsating DC voltage 210 (in FIG. 2a) is provided to the valley-fill circuit 330 through diode 140. Two energy-storage capacitors 280, 282 are provided in the valley-fill circuit 330. These energy-storage capacitors 280, 282 charge in series with a charging current flowing through capacitor 280, diode 284, capacitor 282, and a resistor 286, which limits the magnitude of the charging current. The energy-storage capacitors 280, 282 are sized such that the same voltage, the valley-fill voltage VVF (shown as 235 in FIG. 2a), is produced across each capacitor. The magnitude of the valley-fill voltage VVF is approximately one-half of the peak, VPEAK, of the rectified pulsating DC voltage 210, which is about 200V when VPEAK is 392V. However, the energy-storage capacitors 280, 282 discharge in parallel, with current flowing through diode 288 to allow capacitor 280 to discharge, and through diodes 290 and 292 to allow capacitor 282 to discharge. Thus, a DC bus voltage 220 is formed across the valley-fill circuit 330 as shown in FIG. 2b. 
When the rectified voltage 210 is greater than the valley-fill voltage VVF, i.e. one-half of the peak of the AC mains line voltage, the inverter 160 draws current directly from the AC power supply, through the EMI filter 115 and the rectifier 120, to drive the lamp. When the rectified voltage 210 is less than the valley-fill voltage VVF, then the inverter 160 draws current from the energy-storage capacitors in parallel. This results in the ballast drawing an input current 240 from the AC mains only during a relatively large duration of each line half-cycle centered about the peak of the line voltage, which allows for unwanted harmonics and undesirable total harmonic distortion (THD).
In order to lower the THD, the input current of the ballast should be as sinusoidal as possible (as shown by 250 in FIG. 2c). One approach to making the input current more sinusoidal is to implement power supply 110 as a cat-ear power supply, which ideally draws an input current 260 (shown in FIG. 2d) near the zero crossing of the AC mains input voltage waveform at either the leading edge of each half-cycle, or the trailing edge of each half-cycle, or both. When the current drawn by the cat-ear power supply is added to the inverter current 240, the input current waveform is shaped to be more nearly sinusoidal, such that the input current THD is substantially reduced, and the power factor of the ballast is increased. The cat-ear power supply derives its name from the shape of its input current waveform that “fills in” the current waveform drawn by the ballast from the AC mains around the zero crossings (the shape resembling the ears of a cat). That is, the input current waveform typically rises from zero sinusoidally to a value substantially below peak, then falls sharply to zero, or rises from zero sharply to a value substantially below peak, then falls sinusoidally to zero. The cat-ear power supply typically “steals” power from the line when the back end is not drawing current directly from the line. The cat-ear power supply may be provided with circuitry that “cuts in” and “cuts out” the power supply in response to fixed input voltage levels. Along with helping to reduce THD and improve power factor, the cat-ear power supply also supplies the power necessary to operate the control circuit 190.
A prior art cat-ear power supply 310 is shown in FIG. 3b. The cat-ear power supply 310 is designed with fixed voltage cut-in and cut-out points and will only draw current from the AC mains when the rectified voltage 210 is below a predetermined value. This condition will occur from a predetermined time before a line voltage zero crossing to a predetermined time after the line voltage zero crossing. The cut-out and cut-in voltage points can be adjusted so that the cat-ear power supply 310 draws current during a first interval from a time just after the line voltage zero crossing to a time when the energy storage capacitor in the valley-fill circuit 130 begins drawing charging current from the line, and during a second interval from a time when the valley-fill energy storage capacitor stops drawing charging current from the line until the next line voltage zero crossing.
When the rectified voltage 210 is lower than a predetermined voltage, a charging field effect transistor (FET) 312 conducts to allow charging of energy-storage capacitor 314, which charges toward a voltage VCC. Alternatively, when the rectified line voltage is equal to or greater than the predetermined voltage, then cut-out transistor 318 begins conducting. The collector of the cut-out transistor 318 pulls the cathode of a Zener diode 320 toward VCC, which effectively turns off the charging FET 312. The predetermined cut-in and cut-out voltages are determined by the resistive voltage divider network including resistors 322 and 324, to which the base of the cut-out transistor 318 is connected.
The rate of charge of the capacitor 314 is determined by a resistor 316 in series with the drain of the MOSFET transistor 312. To allow for a substantially piece-wise continuous ballast input current, the value of the current drawn by cat-ear power supply 310 should be substantially the same as the current that will be drawn by the back end of the ballast 100 at the predetermined cut-out and cut-in times. In conjunction with the value of the capacitor 314, resistor 316 can be chosen so that the current drawn will have a desired maximum current that is substantially the same as the current that will be drawn by the back end at the predetermined cut-out and cut-in times and such that the current drawn will substantially match the shape of the AC mains voltage.
However, the current drawn from VCC by the control circuit 190 of ballast 100 is not constant throughout the operation of the ballast. Consequently, the current required to charge capacitor 314 is sometimes smaller, thus the time required to charge capacitor 314 is shorter. Therefore, the current drawn by cat-ear supply 310 sometimes does not reach the desired maximum current at the predetermined cut-out and cut-in times as shown by 360 in FIG. 3c. When the cat-ear supply input current 360 is added to the current 240 drawn by the back end, the resulting ballast input current 370 (shown in FIG. 3d) is not completely sinusoidal, thus contributing to the THD of the ballast.
Additionally, in order to obtain the appropriate shape of the input current waveform, the power dissipated by the resistor may be very large. For example, the power into the cat-ear power supply with the input current 260 (in FIG. 2d) may be approximately four watts each half-cycle. If the maximum power consumption of the control circuit 190 is approximately 0.5 watts, then 3.5 watts must be dissipated in the resistor 316 during each half-cycle. This means that the resistor 316 must be physically large in order to handle the required power dissipation.
Thus, there exists a need for a cat-ear power supply for an electronic ballast that is more efficient and draws the appropriate amount of current when the back end is not drawing current directly from the line in order to reduce the THD of the ballast. Further, there exists a need for an electronic dimming ballast that has the reduced THD of a ballast having an active boost converter, but does not require the large, expensive components of such boost converters.